Communications device and related method that detects symbol timing

ABSTRACT

A communications device includes a signal input for receiving signals such as a binary phase shift keyed (BPSK) communications signal having a repeated preamble bit or symbol pattern. A modem processes the communications signal and includes a demodulator and processor that generates an initial frequency offset estimate and phase error estimate by processing such as with a Fast Fourier Transform (FFT), that detects the repeated preamble pattern for a block of samples within the communications signal, correlates two halves of the block of samples with a plurality of different shifted sequences and determines a maximum correlation value for the shifted sequence that provides the maximum correlation value to establish a symbol timing estimate based on the known timing alignment of this shifted sequence. Radio circuitry is operative with the modem for processing communications data obtained from the communications signal.

FIELD OF THE INVENTION

The present invention relates to communications devices, and moreparticularly, the present invention relates to communications devicesthat improve acquisition estimates of frequency offset and phase error.

BACKGROUND OF THE INVENTION

Some multi-band or other tactical radios operate in the high frequency(HF), very high frequency (VHF), and ultra high frequency (UHF) bands.The frequency range of these multi-band tactical radios is from about 2MHz to about 512 MHz. Next generation radios will probably cover about2.0 to about 2,000 MHz (2.0 GHz) (or higher) to accommodate widerbandwidths, higher data rates and less crowded frequency bands. Severalstandards have been developed for the different frequency bands. For HF,US-MIL-STD-188-110B and US-MIL-STD-188-141B specify waveforms andminimum performance requirements of waveforms and radio equipment, thedisclosures which are incorporated by reference in their entirety.

UHF standards, on the other hand, provide different challenges over the225 to about 512 MHz frequency range, including short-haul line-of-sight(LOS) communication and satellite communications (SATCOM) and cable. UHFwaveforms operate through different weather conditions, foliage andother obstacles making UHF SATCOM an indispensable communications mediumfor many agencies. Different directional antennas can be used to improveantenna gain and improve data rates on the transmit and receive links.This type of communication is typically governed in one example byMIL-STD-188-181B, the disclosure which is incorporated by reference inits entirety. This standard specifies a family of constant andnon-constant amplitude waveforms for use over satellite links.

The joint tactical radio system (JTRS) implements some of thesestandards and has different designs that use oscillators, mixers,switchers, splitters, combiners and power amplifier devices to coverdifferent frequency ranges. The modulation schemes used for these typesof systems can occupy a fixed bandwidth channel at a fixed carrierfrequency or can be frequency-hopped.

These systems use many different types of modulations, including M-aryphase-shift keying (M-PSK) modulation, M-ary quadrature-amplitudemodulation (M-QAM) or modulations with memory, such as continuous phasemodulation (CPM), and are sometimes combined with convolutional or othertype of forward error correction codes. To ensure interoperability,standardized waveforms are often used. These and other systems often usea Binary Phase Shift Keyed (BPSK) waveform for Demand Assigned MultipleAccess (DAMA) communications systems. Some examples are the 117F and F3manpack radios manufactured by Harris Corporation of Melbourne, Fla.Several performance issues were noted in some of these and similarradios as caused by “bad” acquisitions. The receiver modem must acquirethe waveform in each DAMA slot such as corresponding to a time divisionmultiple access (TDMA) slot. If acquisition estimates have excessiveerror, data is lost for the entire slot.

This type of modem often uses a Fast Fourier Transform (FFT) to detectthe waveform and exploit spectral characteristics of a transmittedpreamble. After the modem processes the FFT and detects the preamble,the modem estimates the frequency offset and the phase from complex FFToutput values. It is desirable to improve acquisition estimates of thefrequency offset, phase error and symbol timing to allow betterprocessing and acquisition estimates and enhance communications.

Also, because of the non-random aspect of the preamble, the start ofmessage bit correlation can sometimes false alarm during this non-randompreamble portion of the waveform, i.e., the 110110110110 portion of thepreamble, for example, forming the training sequence. It would beadvantageous if false detections could be reduced in this portion of thepreamble prior to the start of message bits.

SUMMARY OF THE INVENTION

A communications device includes a signal input for receiving digitallymodulated communications signal (such as a binary phase shift keyed(BPSK) signal) having a repeated preamble bit or symbol pattern. A modemprocesses the communications signal and includes a demodulator andprocessor that generates an initial frequency offset estimate and phaseerror estimate by processing a Fast Fourier Transform (FFT) that detectsthe repeated preamble bit or symbol pattern for a block of sampleswithin the communications signal, correlates two halves of the block ofsamples with a plurality of different shifted sequences and determines amaximum correlation value based on the shifted sequence that providesthe maximum correlation value to establish a symbol timing estimatebased on the known timing alignment of the shifted sequence providingthe maximum correlation value. Radio circuitry is operative with themodem for processing communications data obtained from thecommunications signal.

The modem is operative for rotating the block of samples by the negativeof an initial frequency offset estimate, in order to compensate for thereceived frequency offset. It can calculate the magnitude-squared of thesum of the two halves of the complex correlation outputs for theplurality of different shifted sequences to obtain a symbol timingestimate based on the maximum magnitude-squared sum value. The repeatedpreamble pattern can be formed as a training sequence and the initialsignal detection spans 256 samples containing 64 symbols sampled at foursamples per symbol in one embodiment.

BRIEF DESCRIPTION OF THE DRAWINGS

Other objects, features and advantages of the present invention willbecome apparent from the detailed description of the invention whichfollows, when considered in light of the accompanying drawings in which:

FIG. 1 is a block diagram of a transceiver modem that can be used inaccordance with a non-limiting example of the present invention forimproving estimates of frequency offset, phase error and symbol timingand better detecting the start of message sequence.

FIG. 2 shows a repeated pattern in an initial communications signaltransmission and showing a start of message (SOM) sequence.

FIG. 3 is a high-level flowchart showing an example of a method used forimproving acquisition estimates of frequency offset, phase error andsymbol timing in accordance with a non-limiting example of the presentinvention.

FIG. 4 is another high-level flowchart similar to the flowchart shown inFIG. 3, and showing when the symbol timing estimate is obtained relativeto when the frequency and phase estimates are obtained.

FIGS. 5-7 are graphs showing models of estimated frequency offsethistograms for different signal-to-noise (Eb/No) ratios and frequencyoffsets in accordance with a non-limiting example of the presentinvention.

FIG. 8 is a graph showing a model of the bit error rate for differentradios, including the radio using the iterative technique in accordancewith a non-limiting example of the present invention and a legacy radio.

FIG. 9 is a graph showing a model of the probability of good acquisitionfor a radio using the iterative technique in accordance with anon-limiting example of the present invention as compared to an existinglegacy radio.

FIG. 10 is a graph showing a model of the probability of falseacquisition and comparing legacy radios and an initial performance witha radio using the iterative technique using the system and method inaccordance with a non-limiting example of the present invention.

FIG. 11 is a block diagram showing a correlator with a number ofdifferent correlator modules for detecting a start of message sequencein accordance with a non-limiting example of the present invention.

FIG. 12 is a graph showing a model of the correlation results during afalse acquisition using four correlators.

FIG. 13 is a graph showing a model of the correlation results for thefour correlators in a good acquisition in accordance with a non-limitingexample of the present invention.

FIG. 14 is a block diagram of an example of a communications system thatcan be used in accordance with a non-limiting example of the presentinvention.

FIG. 15 is a high-level block diagram showing basic components that canbe used in accordance with a non-limiting example of the presentinvention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Different embodiments will now be described more fully hereinafter withreference to the accompanying drawings, in which preferred embodimentsare shown. Many different forms can be set forth and describedembodiments should not be construed as limited to the embodiments setforth herein. Rather, these embodiments are provided so that thisdisclosure will be thorough and complete, and will fully convey thescope to those skilled in the art.

It should be appreciated by one skilled in the art that the approach tobe described is not limited for use with any particular communicationstandard (wireless or otherwise) and can be adapted for use withnumerous wireless (or wired) communications standards such as EnhancedData rates for GSM Evolution (EDGE), General Packet Radio Service (GPRS)or Enhanced GPRS (EGPRS), extended data rate Bluetooth, Wideband CodeDivision Multiple Access (WCDMA), Wireless LAN (WLAN), Ultra Wideband(UWB), coaxial cable, radar, optical, etc. Further, the invention is notlimited for use with a specific PHY or radio type but is applicable toother compatible technologies as well.

Throughout this description, the term communications device is definedas any apparatus or mechanism adapted to transmit, receive or transmitand receive data through a medium. The communications device may beadapted to communicate over any suitable medium such as RF, wireless,infrared, optical, wired, microwave, etc. In the case of wirelesscommunications, the communications device may comprise an RFtransmitter, RF receiver, RF transceiver or any combination thereof.Wireless communication involves: radio frequency communication;microwave communication, for example long-range line-of-sight via highlydirectional antennas, or short-range communication; and/or infrared (IR)short-range communication. Applications may involve point-to-pointcommunication, point-to-multipoint communication, broadcasting, cellularnetworks and other wireless networks.

As will be appreciated by those skilled in the art, a method, dataprocessing system, or computer program product can embody differentexamples in accordance with a non-limiting example of the presentinvention. Accordingly, these portions may take the form of an entirelyhardware embodiment, an entirely software embodiment, or an embodimentcombining software and hardware aspects. Furthermore, portions may be acomputer program product on a computer-usable storage medium havingcomputer readable program code on the medium. Any suitable computerreadable medium may be utilized including, but not limited to, staticand dynamic storage devices, hard disks, optical storage devices, andmagnetic storage devices.

The description as presented below can apply with reference to flowchartillustrations of methods, systems, and computer program productsaccording to an embodiment of the invention. It will be understood thatblocks of the illustrations, and combinations of blocks in theillustrations, can be implemented by computer program instructions.These computer program instructions may be provided to a processor of ageneral purpose computer, special purpose computer, or otherprogrammable data processing apparatus to produce a machine, such thatthe instructions, which execute via the processor of the computer orother programmable data processing apparatus, implement the functionsspecified in the block or blocks.

These computer program instructions may also be stored in acomputer-readable memory that can direct a computer or otherprogrammable data processing apparatus to function in a particularmanner, such that the instructions stored in the computer-readablememory result in an article of manufacture including instructions whichimplement the function specified in the flowchart block or blocks. Thecomputer program instructions may also be loaded onto a computer orother programmable data processing apparatus to cause a series ofoperational steps to be performed on the computer or other programmableapparatus to produce a computer implemented process such that theinstructions which execute on the computer or other programmableapparatus provide steps for implementing the functions specified in theflowchart block or blocks.

The communications system and device and related method, in accordancewith a non-limiting example of the present invention, involves aniterative approach to refine frequency offset and phase error estimatesand provides a highly accurate symbol timing estimate.

Some current modems for communications systems and devices, especiallythose processing a BPSK waveform, have implemented an approach that usesthe FFT to detect the waveform by exploiting the spectralcharacteristics of the transmitted preamble. After the FFT detects thepreamble, the modem estimates the frequency offset and phase error fromcomplex FFT output values. Note that other techniques (such as slidingblock correlators using a single reference sequence rotated by variousfrequency offset hypothesis, where block size is equal to FFT size) canbe used instead of FFT approach but may be more computational intensiveto implement.

An initial transmission typically has a repeated pattern such as 011,followed by a unique start of message (SOM) sequence. The FFT basedprocessing in the modem detects the repeated pattern of the 011 sequenceand generates an initial frequency offset and phase error estimate. TheFFT in this case spans 256 samples containing 64 BPSK symbols sampled atfour samples per symbol. The system, device and method in accordancewith a non-limiting example of the present invention processes thisblock of 256 samples and the initial frequency offset estimate in aniterative fashion to improve the acquisition estimates of frequencyoffset and phase error and provide a symbol timing estimate. It alsobetter detects the start of message sequence using correlators. Notethat the block size can be increased in an effort to further improve theinitial acquisition processing (for example, 512 samples instead of256).

An example of the steps that can be used, in accordance with anon-limiting example of the iterative technique, is set forth below asexplained for a BPSK waveform.

Step 1: The 256 samples are rotated by the negative of the initialfrequency offset estimate.

Step 2: A complex correlation is performed (in two halves, 128 sampleseach) with 12 different BPSK 011 training sequence alignments (4 samplesper symbol) as follows:

-   -   000011111111000011111111000011111111 . . .        00011111111000011111111000011111111 . . .        00111111110000111111110000111111110000 . . .        0111111110000111111110000111111110000 . . .        111111110000111111110000111111110000 . . .        111111100001111111100001111111100001111 . . .        1111110000111111110000111111110000111111 . . .        11111000011111111000011111111000011111111 . . .        111100001111111100001111111100001111111100 . . .        111000011111111000011111111000011111111000 . . .        110000111111110000111111110000111111110000 . . .        10000111111110000111111110000111111110000 . . . .        Note that each training sequence alignment will be extended to        be 256 samples long to match the block size of step 1.

Step 3: The magnitude-squared of the sum of the two halves of eachcorrelator output is calculated and the alignment giving the maximumvalue is selected Note that only the training sequence alignmentyielding the maximum value will be used in steps 4-7.

Step 4: A frequency offset adjustment is calculated by performing thecomplex conjugate dot product between the first one-half complexcorrelator output and the second half complex correlator output.

Step 5: The frequency offset estimate is updated to the previousestimate plus the new adjustment.

Step 6: The initial 256 samples used by FFT processing are rotated bythe negative of the updated frequency offset estimate.

Step 7: A complex correlation is performed (in two halves, 128 sampleseach) with the training sequence alignment that gave the maximum value.

Step 8: Steps 4-7 are repeated a number of times (iteratively). In onenon-limiting example, three iterations occur.

Step 9: The final updated frequency estimate is obtained after theiterative process, the selected training sequence alignment accuratelyidentifies the proper symbol alignment and the final correlation valuesprovide the phase error estimate to give a more refined (or accurate)frequency offset and phase error.

FIG. 1 is a high-level block diagram of a communications device 20, forexample, a radio, which includes, as a non-limiting example, a BPSKmodem that can be used for processing a BPSK waveform for DAMAoperation. In this example, demand assigned multiple access (DAMA)technology assigns a bandwidth to clients that do not require use of thebandwidth constantly. Typically, DAMA communications systems assigncommunications channels or circuits based on requests from userterminals to network control systems. When the circuit is not in use,channels can be returned for reuse by others. This technology is oftenused in satellite channels on a per request basis and increases theamount of users in a pool of channels available for use by any stationin a network.

DAMA technology is often used in association with a BPSK waveformbecause binary phase shift keying (BPSK) is a simple and robust digitalmodulation technique. BPSK uses two phases that are separated by 180degrees (thus it can also be termed 2-PSK). This type of modulation ismore robust than higher-order PSK modulations (i.e. 4-PSK, 8-PSK, etc)and serious distortions would have to occur to force a demodulator toreach an incorrect decision.

It should be understood that the technique as described can be used withany type of modem that processes M-PSK, M-QAM and CPM waveforms andparticularly a BPSK waveform that has known repeated data in thepreamble to provide a technique for not only obtaining the symboltiming, but also improving the acquisition estimates of the frequencyoffset and phase error. Also, as noted before, the modem uses the FFT todetect the waveform by exploiting the spectral characteristics of atransmitted preamble.

As shown in FIG. 1, the modem 21 is incorporated as part of atransceiver modem processor for BPSK modulation and demodulation andincludes a modulator 22 and demodulator 24. The modulator 22 receivesdata and includes all necessary processing functions with appropriatemodules and circuits as required for operation. A clock signal isreceived from a system clock as illustrated. The demodulator 24 includesnecessary functional modules or circuits, including those for adetector, demodulator, decoder, data decompressor, CRC checker, andfilter as non-limiting examples of the type of circuits or modules thatcan be part of the modem and its processor. The modem 21 can beincorporated within a digital signal processor or field programmablegate array, including a software defined radio, or it can be part of ahard-wired transceiver or discrete modem radio circuit. The modemincludes any necessary processor modules 24 b. The switch 26 allowstransmit and receive functions to pass through other radio circuits 28into and from the antenna 30.

Appropriate amplifiers 32,34 can be positioned between the modulator 22,switch 26 and demodulator 24. Correlator modules 24 a, as will beexplained in further detail relative to FIG. 11, detect the start ofmessage sequence in a better manner with reduced false detects,providing a technique for reducing start of message false detects. Thecorrelators using multiple hypothesis correlation processing as a “quadcorrelator.” Due to the non-random aspect of the preamble, the start ofmessage bit correlation can sometimes false alarm during this non-randompreamble portion of the waveform. The correlator modules can reduce thefalse detections in the (110110 . . . ) portion of the preamble prior tothe start of message bits. Once the waveform preamble is acquired andfrequency phase and timing offset computed, it is possible to start thecorrelator modules while searching for the start of message bits and theother plurality of correlators correlating for three different phases ofthe 110 pattern, as will be explained in detail below.

It should be understood that different techniques can be used forgenerating BPSK signals, including the use of standard lattice ringmodulators or balanced modulators. Demodulation can occur using abalanced modulator, such as a diode ring or lattice modulator. Thesemodules or circuit functions could be incorporated within appropriateDSP or FPGA circuits that include the modem. As an example, whendemodulating BPSK signals, a carrier with a correct frequency and phaserelationship can be applied to a balanced modulator along with the BPSKsignal in an appropriate carrier recovery circuit. Some type of bandpassfilter could ensure that only the desired BPSK signal is passed. Thissignal could be squared or multiplied by itself in a balanced modulatoror analog multiplier by applying the same signal to both inputs.Squaring could remove the 180 degree phase shifts resulting in an outputthat is twice the input signal frequency. The signal could be passedthrough a bandpass filter with the resulting signal applied to the phasedetector of a phase locked loop (PLL). A voltage controlled oscillatorcould track any carrier frequency shifts. A correct frequency and phaserelationship can be obtained as the result, and the carrier is appliedto the balanced modulator/demodulator along with a BPSK signal in onenon-limiting example.

FIG. 2 shows a bit pattern for an initial transmission of a BPSKcommunications signal. This pattern shows the repeated pattern of 011 asa training sequence in one non-limiting example at 35, followed by aunique start of message sequence, illustrated with a portion of the bitpattern sequence at 36. The FFT based processing detects this repeated011 bit pattern sequence as a training sequence (or preamble) andgenerates an initial frequency offset and phase error estimate. The FFTspans 256 samples which contain 64 BPSK symbols sampled at four samplesper symbol in this non-limiting example. As noted before, the receivermodem processes this block of 256 samples and the initial frequencyoffset estimate in an iterative fashion to improve acquisition estimatesof frequency offset and phase error. It can also obtain symbol timing ina previous portion of the technique.

FIG. 3 is a high-level flowchart in accordance with a non-limitingexample of the present invention. The samples are rotated by thenegative of the initial frequency offset estimate that is obtainedthrough the FFT (block 40). The FFT based processing looks for thefrequency components with specific spectral properties and determineswhen the signal is present. In the following description, the term modemis used and can apply to the modem functions of a communications deviceand any associated circuitry.

A complex correlation is next performed in two halves with differentBPSK training sequence alignments (block 42). As noted before in theprevious step-by-step description, twelve different BPSK trainingsequence alignments are used in this example. Any number of differentBPSK sequences, however, could possibly be used, depending on end-useand device processing requirements (such as the number of samples persymbol). It should be understood that the phasing of the 011 portions inthe data are initially not known. Because there are 256 samples, the bitsequence is broken into two halves, with 128 samples followed by 128samples. These samples are correlated against a shifted and repeated 011BPSK symbol pattern. For example, if four samples per symbol are used,the 011 BPSK symbol pattern expands to 000011111111 samples and thisbase pattern is repeated to generate 256 samples. To generate the secondsequence, the base pattern is circularly shifted left by 1. The thirdpattern is the base pattern circularly shifted left by 2, the fourthpattern is the base pattern circularly shifted left by 3 and so forth.Once the twelve patterns are generated, the processor can then correlatethe received 256 samples with the 12 training sequence alignmentpatterns, where each correlation is over 128 samples.

At this time (block 44), the sum of the magnitude squared of the twohalves is calculated using the correlation outputs of each of the 12training sequence alignment patterns. The modem receive processing looksat the 12 magnitude squared values and the largest value corresponds tothe best symbol timing alignment. Thus, this technique can determinesymbol timing alignment down to a sample. Prior to this step, the modemdoes not know where symbol boundaries begin and end.

At this time, the modem calculates the frequency offset adjustment byperforming a complex conjugate dot product (block 46) between the outputof the two 128 sample correlations for the sequence alignment with thelargest magnitude squared value. Ideally, if the first estimate comingfrom the FFT was correct, this would be a 0. This often would not occurand as a result, a delta (corresponding to the change) in the frequencyoffset is obtained. The frequency offset estimate is updated (block 48)as based on the initial value with the new estimated value. A loop backoccurs and the initial data is rotated by the negative of the newfrequency offset value. The initial FFT values are rotated and a complexcorrelation performed with only the shifted training sequence alignmentpattern that gave the largest magnitude squared value (block 50).Instead of performing the correlation for all 12 sequences, the modemperforms a correlation for one sequence in which the timing alignment isknown as has been determined in the previous portion of the process. Themodem the first time through the process obtains the symbol timingalignment from the sequence providing the maximum value. On thesubsequent rotation, the loop occurs.

The frequency offset adjustment is repeated (block 52) and the frequencyoffset estimate is updated with the complex correlation an “N” number oftimes. “N” can vary, but in one non-limiting example, is three. Thefinal updated frequency estimate is used and the symbol alignment isidentified and the phase error estimate completed (block 54). Thus, theselected correlator gives the proper symbol timing alignment and thefinal correlator values for the phase error estimate.

FIG. 4 is another high-level flowchart showing the basic sequences ofoperation in accordance with a non-limiting example of the presentinvention. The samples are received (block 60) and the initial FET andfrequency estimate are established (block 62). The frequencycompensation is determined (block 64). The twelve sequence correlationsoccur (block 66) and the proper symbol timing alignment is picked (block68), resulting in the symbol timing estimate. A new frequency estimateis established (block 70). The frequency estimate is updated (block 72)and the frequency compensation, by rotating by the negative of thefrequency estimate occurs (block 74). The best training sequencealignment is used for the correlation (block 76) and the new frequencyestimate established (block 78). This process is repeated a plurality oftimes for N iterations as seen by the loop back. The final frequency andphase estimates are established (block 80). It should be understood thatthis technique can be used with any type of modem that sends a knownsequence as a preamble (or training sequence) to improve the frequencyestimate and phase estimate.

It is possible for the line drawn from the frequency compensation (block64) to come from after the received samples at block 60 into thefrequency compensation at block 74. This process and choice can dependon how the frequency estimate is maintained and the delta compensated.It is possible to establish an initial frequency estimate from theacquisition and keep updating a delta frequency estimate from thatestimation. It is also possible to maintain a total frequency offset andpull the received samples every time.

FIGS. 5-7 are graphs of models showing the estimated frequency offsethistogram for different signal-to-noise ratios (Eb/No) and differentfrequency offsets in hertz (Hz) and comparing an initial estimate withthe iterative technique in accordance with a non-limiting example of thepresent invention. The frequency offset is shown on the horizontal Xaxis and the number of occurrences are shown on the vertical Y axis.

FIG. 5 shows the frequency offset estimate relative to the number ofoccurrences when the Eb/No is 2 and no frequency offset is inserted intosystem.

FIG. 6 shows the estimated frequency offset histogram when the Eb/No is10 and no frequency offset is inserted into system.

FIG. 7 shows the estimated frequency offset for a 100 Hz offset insertedinto system when the Eb/No is 2.

When no offset is inserted, ideally only an impulse centered at zerowould be observed. When an offset is inserted into system, impulseshifts to frequency offset value. Noise causes the FFT and the newtechnique to differ from the ideal impulse centered at 0 (for FIGS. 5and 6) or 100 Hz (for FIG. 7). It is evident from these results that atlow signal-to-noise ratios, the new technique reduces the frequencyerror computed by the FFT process.

FIG. 8 shows the bit error rate (BER) performance for an existing legacyradio, the performance when only the FFT approach is used (labeled“Initial Performance”) and the performance using a non-limiting exampleof the present invention (labeled “After Iterative Technique”). Clearly,a significant improvement is observed when present invention is used inreceive processing.

FIG. 9 shows the probability of good acquisition for the same 3 systemsof FIG. 8. Present invention performs as well as “Legacy Radio” whileproviding superior BER performance (FIG. 8).

FIG. 10 shows that the probability of a false acquisition for theiterative technique compared to the performance of initial system.Again, significant performance benefits from present invention can beobserved.

For the case of BPSK, the communications device and modem provides forfast complex correlation processing when using the iterative techniqueas described because the reference patterns are purely real (i.e. thecomputational complexity of correlations is half of true complexcorrelations). Additionally, for all modulation types, computationalcomplexity is reduced further by using only one training sequencealignment pattern in iterative process. It should be understood that notonly does the modem use the 12 training sequence alignment patterncorrelations to obtain good symbol timing, but also through furtherprocessing the modem and its iterative processing obtains the enhancedfrequency and phase estimate. The iterative technique as described canbe applied to any waveform (and modulation) that has a known repeatedpreamble. The first portion of the algorithm which is the firstiteration of the iterative technique obtains a better alignment for thesymbol timing. Then, a better frequency estimate and phase estimate areobtained as the iterative technique proceeds.

Referring now to FIG. 11, the details of the correlator modules 24 ashown in FIG. 1 are illustrated. For the example case of BPSK with arepeated training sequence of 011, four correlators are shown. The firstcorrelator is as a start of message correlator 90 that is operativeusing a start of message bit sequence derived from a Legendre Polynomial(LPN). Three training correlators 91, 92, 93 are operative with thethree different phases of the 011 sequence, e.g., the 110, 011, and 101phase as illustrated. The processor module 24 b shown in FIG. 1 is alsooperative with the correlators 90, 91, 92, 93. Note that one sample persymbol processing is used for this portion of modem processing, in anon-limiting example.

The correlators 90, 91, 92, 93 (i.e. quad correlators) together reducethe false detection of the start of message sequence. As noted before,the modem must acquire the waveform in each DAMA slot. Because of thenon-random aspect of the preamble, the start of message bit correlationcan sometimes false alarm during this non-random preamble portion of thewaveform, i.e., 110110110110 (see FIG. 12). The quad correlator reducesthe false detections in the 110110 . . . portion of the preamble priorto the start of message bits as shown in FIG. 13.

As noted before, some legacy radios use the FFT to detect the waveformby exploiting the spectral characteristics of the transmitted preamble.After the FFT detects the preamble, the modem estimates the frequencyoffset, phase error and symbol timing. To detect the start of message, astart of message bit correlator is executed until the correlator outputvalue exceeds a threshold. In these legacy radios, some operability isobtained with adequate results, but better results could be obtained.

The start of message false detect technique, in accordance with anon-limiting example of the present invention, acquires the waveformpreamble and the frequency offset, phase error and symbol timing arecomputed as noted before. The four correlators are started. The firstcorrelator as the start of message correlator searches for the start ofmessage bits. The other three correlators correlate for the threedifferent phases of the 110 pattern. The start of message correlatoroutput 90 must be larger than any one of the three correlators 91, 92,93 looking for the different phases of the 110 pattern. The processormodule 24 b provides a new state machine for acquisition with thecorrelators. It should be understood that the processor module of themodem can incorporate the different correlators as part of a digitalsignal processor or field programmable gate array or other processingmodules or circuitry. The processor module takes the results from thecorrelator and processes them for the start of message detect. It shouldbe understood that the start of message should be greater than the startof message threshold and all three correlators should have less than astart of message threshold to correspond to a good acquisition. Thus,the correlator module 24 a avoids falsing during the non-random portionof a preamble when searching for the start of message bits.

It should be understood that the correlator modules are advantageousbecause the 011 or similar pattern often is a poor pattern from acorrelation properties perspective and this pattern makes it difficultto correlate for the start of message, especially if noise is involved.The quad correlator allows a more reliable detection of the start ofmessage because the three different phases of the 011 are looked forwithin the correlators. In the first portion of the technique describedrelative to FIGS. 1-10, the four samples for every symbol are processed.In a non-limiting example of the present invention, the start of messagefalse detect technique described by FIG. 11 shows modem processingperformed on a one sample per symbol basis with four differentcorrelators running: the start of message correlator and the threephases of the 011 pattern.

The first correlator as the start of message correlator 90 uses bitsobtained from a Legendre Polynomial (LPN). Note that this bit patterncan be found in the US MIL-STD-188-181B specification.

FIGS. 12 and 13 show respective models for the correlation results witha false acquisition shown in FIG. 12 and a good acquisition shown inFIG. 13. The signal-to-noise ratio is 2 dB in each graph with the symbolnumber shown on the horizontal X axis and the correlation result shownon the vertical Y axis. The four correlation results for the LPN as thestart of message correlator is shown with the three correlators astraining correlators for the 110, 011 and 101 correlator. The thresholdof 600 is used in the model of FIG. 13.

As shown in FIG. 12, the SON correlator has a false peak during thetraining sequence. Modem processing that only correlates for the SONsequence would have falsely detected the SON pattern and outputerroneous data for that DAMA slot.

In the graph shown in FIG. 13, there is a good acquisition with thethree training correlators coming up and ramping back down again. Thepeak occurs at the end after the three correlators slide into the startof message bit pattern. The phase can be irrelevant at the end and apositive or negative peak can occur.

For purposes of description, some background information on coding,interleaving, and an exemplary wireless, mobile radio communicationssystem that includes ad-hoc capability and can be modified for use isset forth. This example of a communications system that can be used andmodified for use with the present invention is now set forth with regardto FIGS. 14 and 15.

An example of a radio that could be used with such system and method isa Falcon™ III radio manufactured and sold by Harris Corporation ofMelbourne, Fla. This type of radio can support carrier frequencies form30 MHz up to 2 GHz, including L-band SATCOM and MANET. The waveforms canprovide secure IP data networking. It should be understood thatdifferent radios can be used, including software defined radios that canbe typically implemented with relatively standard processor and hardwarecomponents. One particular class of software radio is the Joint TacticalRadio (JTR), which includes relatively standard radio and processinghardware along with any appropriate waveform software modules toimplement the communication waveforms a radio will use. JTR radios alsouse operating system software that conforms with the softwarecommunications architecture (SCA) specification (seewww.jtrs.saalt.mil), which is hereby incorporated by reference in itsentirety. The SCA is an open architecture framework that specifies howhardware and software components are to interoperate so that differentmanufacturers and developers can readily integrate the respectivecomponents into a single device.

The Joint Tactical Radio System (JTRS) Software Component Architecture(SCA) defines a set of interfaces and protocols, often based on theCommon Object Request Broker Architecture (CORBA), for implementing aSoftware Defined Radio (SDR). In part, JTRS and its SCA are used with afamily of software re-programmable radios. As such, the SCA is aspecific set of rules, methods, and design criteria for implementingsoftware re-programmable digital radios.

The JTRS SCA specification is published by the JTRS Joint Program Office(JPO). The JTRS SCA has been structured to provide for portability ofapplications software between different JTRS SCA implementations,leverage commercial standards to reduce development cost, reducedevelopment time of new waveforms through the ability to reuse designmodules, and build on evolving commercial frameworks and architectures.

The JTRS SCA is not a system specification, as it is intended to beimplementation independent, but a set of rules that constrain the designof systems to achieve desired JTRS objectives. The software framework ofthe JTRS SCA defines the Operating Environment (OE) and specifies theservices and interfaces that applications use from that environment. TheSCA OE comprises a Core Framework (CF), a CORBA middleware, and anOperating System (OS) based on the Portable Operating System Interface(POSIX) with associated board support packages. The JTRS SCA alsoprovides a building block structure (defined in the API Supplement) fordefining application programming interfaces (APIs) between applicationsoftware components.

The JTRS SCA Core Framework (CF) is an architectural concept definingthe essential, “core” set of open software Interfaces and Profiles thatprovide for the deployment, management, interconnection, andintercommunication of software application components in embedded,distributed-computing communication systems. Interfaces may be definedin the JTRS SCA Specification. However, developers may implement some ofthem, some may be implemented by non-core applications (i.e., waveforms,etc.), and some may be implemented by hardware device providers.

For purposes of description only, a brief description of an example of acommunications system that includes communications devices thatincorporate the filter in accordance with a non-limiting example, isdescribed relative to a non-limiting example shown in FIG. 14. Thishigh-level block diagram of a communications system includes a basestation segment and wireless message terminals that could be modifiedfor use with the present invention. The base station segment includes aVHF radio 160 and HF radio 162 that communicate and transmit voice ordata over a wireless link to a VHF net 164 or HF net 166, each whichinclude a number of respective VHF radios 168 and HF radios 170, andpersonal computer workstations 172 connected to the radios 168,170.Ad-hoc communication networks 173 are interoperative with the variouscomponents as illustrated. The entire network can be ad-hoc and includesource, destination and neighboring mobile nodes. Thus, it should beunderstood that the HF or VHF networks include HF and VHF net segmentsthat are infrastructure-less and operative as the ad-hoc communicationsnetwork. Although UHF and higher frequency radios and net segments arenot illustrated, these could be included.

The radio can include a demodulator circuit 162 a and appropriateconvolutional encoder circuit 162 b, block interleaver 162 c, datarandomizer circuit 162 d, data and framing circuit 162 e, modulationcircuit 162 f, matched filter circuit 162 g, block or symbol equalizercircuit 162 h with an appropriate clamping device, deinterleaver anddecoder circuit 162 i modem 162 j, and power adaptation circuit 162 k asnon-limiting examples. A vocoder circuit 162 l can incorporate thedecode and encode functions and a conversion unit could be a combinationof the various circuits as described or a separate circuit. A clockcircuit 162 m can establish the physical clock time and through secondorder calculations as described below, a virtual clock time. The networkcan have an overall network clock time. These and other circuits operateto perform any functions necessary for the present invention, as well asother functions suggested by those skilled in the art. Other illustratedradios, including all VHF (or UHF) and higher frequency mobile radiosand transmitting and receiving stations can have similar functionalcircuits. Radios could range from 30 MHz to about 2 GHz as non-limitingexamples.

The base station segment includes a landline connection to a publicswitched telephone network (PSTN) 180, which connects to a PABX 182. Asatellite interface 184, such as a satellite ground station, connects tothe PABX 182, which connects to processors forming wireless gateways 186a, 186 b. These interconnect to the VHF radio 160 or HF radio 162,respectively. The processors are connected through a local area networkto the PABX 182 and e-mail clients 190. The radios include appropriatesignal generators and modulators.

An Ethernet/TCP-IP local area network could operate as a “radio” mailserver. E-mail messages could be sent over radio links and local airnetworks using STANAG-5066 as second-generation protocols/waveforms, thedisclosure which is hereby incorporated by reference in its entiretyand, of course, preferably with the third-generation interoperabilitystandard: STANAG-4538, the disclosure which is hereby incorporated byreference in its entirety. An interoperability standard FED-STD-1052,the disclosure which is hereby incorporated by reference in itsentirety, could be used with legacy wireless devices. Examples ofequipment that can be used in the present invention include differentwireless gateway and radios manufactured by Harris Corporation ofMelbourne, Fla. This equipment could include RF5800, 5022, 7210, 5710,5285 and PPC 117 and 138 series equipment and devices as non-limitingexamples.

These systems can be operable with RF-5710A high-frequency (HF) modemsand with the NATO standard known as STANAG 4539, the disclosure which ishereby incorporated by reference in its entirety, which provides fortransmission of long distance radio at rates up to 9,600 bps. Inaddition to modem technology, those systems can use wireless emailproducts that use a suite of data-link protocols designed and perfectedfor stressed tactical channels, such as the STANAG 4538 or STANAG 5066,the disclosures which are hereby incorporated by reference in theirentirety. It is also possible to use a fixed, non-adaptive data rate ashigh as 19,200 bps with a radio set to ISB mode and an HF modem set to afixed data rate. It is possible to use code combining techniques andARQ.

A communications system that incorporates communications devices can beused in accordance with non-limiting examples of the present inventionand is shown in FIG. 14. A transmitter is shown at 191 and includesbasic functional circuit components or modules, including a forwarderror correction encoder 192 a that includes a puncturing module, whichcould be integral to the encoder or a separate module. The decoder 192 aand its puncturing module includes a function for repeating as will beexplained below. Encoded data is interleaved at an interleaver 192 b,for example, a block interleaver, and in many cases modulated atmodulator 192 c. This modulator can map the communications data intodifferent symbols based on a specific mapping algorithm to form acommunications signal. For example, it could form Minimum Shift Keyingor Gaussian Minimum Shift Keying (MSK or GMSK) symbols. Other types ofmodulation could be used in accordance with non-limiting examples of thepresent invention. Up-conversion and filtering occurs at an up-converterand filter 192 d, which could be formed as an integrated module orseparate modules. Communications signals are transmitted, for example,wirelessly to receiver 193.

At the receiver 193, down conversion and filtering occurs at a downconverter and filter 194 a, which could be integrated or separatemodules. The signal is demodulated at demodulator 194 b anddeinterleaved at deinterleaver 194 c. The deinterleaved data (i.e. bitsoft decisions) is decoded and depunctured (for punctured codes),combined (for repeated codes) and passed through (for standard codes) atdecoder 194 d, which could include a separate or integrated depuncturingmodule. The system, apparatus and method can use different modules anddifferent functions. These components as described could typically becontained within one transceiver.

It should be understood, in one non-limiting aspect of the presentinvention, a rate 1/2, K=7 convolutional code can be used as an industrystandard code for forward error correction (FEC) during encoding. Forpurposes of understanding, a more detailed description of basiccomponents now follows. A convolutional code is an error-correctingcode, and usually has three parameters (n, k, m) with n equal to thenumber of output bits, k equal to the number of input bits, and m equalto the number of memory registers, in one non-limiting example. Thequantity k/n could be called the code rate with this definition and is ameasure of the efficiency of the code. K and n parameters can range from1 to 8, m can range from 2 to 10, and the code rate can range from 1/8to 7/8 in non-limiting examples. Sometimes convolutional code chips arespecified by parameters (n, k, L) with L equal to the constraint lengthof the code as L=k (m−1). Thus, the constraint length can represent thenumber of bits in an encoder memory that would affect the generation ofn output bits. Sometimes the letters may be switched depending on thedefinitions used.

The transformation of the encoded data is a function of the informationsymbols and the constraint length of the code. Single bit input codescan produce punctured codes that give different code rates. For example,when a rate 1/2 code is used, the transmission of a subset of the outputbits of the encoder can convert the rate 1/2 code into a rate 2/3 code.Thus, one hardware circuit or module can produce codes of differentrates. Punctured codes allow rates to be changed dynamically throughsoftware or hardware depending on channel conditions, such as rain orother channel impairing conditions.

An encoder for a convolutional code typically uses a look-up table forencoding, which usually includes an input bit as well as a number ofprevious input bits (known as the state of the encoder), the table valuebeing the output bit or bits of the encoder. It is possible to view theencoder function as a state diagram, a tree diagram or a trellisdiagram.

Decoding systems for convolutional codes can use 1) sequential decoding,or 2) maximum likelihood decoding, also referred to as Viterbi decoding,which typically is more desirable. Sequential decoding allows bothforward and backward movement through the trellis. Viterbi decoding asmaximum likelihood decoding examines a receive sequence of given length,computes a metric for each path, and makes a decision based on themetric.

Puncturing convolutional codes is a common practice in some systems andis used in accordance with non-limiting examples of the presentinvention. It should be understood that in some examples a puncturedconvolutional code is a higher rate code obtained by the periodicelimination of specific code bits from the output of a low rate encoder.Punctured convolutional code performance can be degraded compared withoriginal codes, but typically the coding rate increases.

Some of the basic components that could be used as non-limiting examplesof the present invention include a transmitter that incorporates aconvolutional encoder, which encodes a sequence of binary input vectorsto produce the sequence of binary output vectors and can be definedusing a trellis structure. An interleaver, for example, a blockinterleaver, can permute the bits of the output vectors. The interleaveddata would also be modulated at the transmitter (by mapping to transmitsymbols) and transmitted. At a receiver, a demodulator demodulates thesignal.

A block deinterleaver recovers the bits that were interleaved. A Viterbidecoder could decode the deinterleaved bit soft decisions to producebinary output data.

Often a Viterbi forward error correction module or core is used thatwould include a convolutional encoder and Viterbi decoder as part of aradio transceiver as described above. For example if the constraintlength of the convolutional code is 7, the encoder and Viterbi decodercould support selectable code rates of 1/2, 2/3, 3/4, 4/5, 5/6, 6/7, 7/8using industry standard puncturing algorithms.

Different design and block systems parameters could include theconstraint length as a number of input bits over which the convolutionalcode is computed, and a convolutional code rate as the ratio of theinput to output bits for the convolutional encoder. The puncturing ratecould include a ratio of input to output bits for the convolutionalencoder using the puncturing process, for example, derived from a rate1/2 code.

The Viterbi decoder parameters could include the convolutional code rateas a ratio of input to output bits for the convolutional encoder. Thepuncture rate could be the ratio of input to output bits for theconvolutional encoder using a puncturing process and can be derived froma rate 1/2 mother code. The input bits could be the number of processingbits for the decoder. The Viterbi input width could be the width ofinput data (i.e. soft decisions) to the Viterbi decoder. A metricregister length could be the width of registers storing the metrics. Atrace back depth could be the length of path required by the Viterbidecoder to compute the most likely decoded bit value. The size of thememory storing the path metrics information for the decoding processcould be the memory size. In some instances, a Viterbi decoder couldinclude a First-In/First-Out (FIFO) buffer between depuncture andViterbi function blocks or modules. The Viterbi output width could bethe width of input data to the Viterbi decoder.

The encoder could include a puncturing block circuit or module as notedabove. Usually a convolutional encoder may have a constraint length of 7and take the form of a shift register with a number of elements, forexample, 6. One bit can be input for each clock cycle. Thus, the outputbits could be defined by a combination of shift register elements usinga standard generator code and be concatenated to form an encoded outputsequence. There could be a serial or parallel byte data interface at theinput. The output width could be programmable depending on the puncturedcode rate of the application.

A Viterbi decoder in non-limiting examples could divide the input datastream into blocks, and estimate the most likely data sequence. Eachdecoded data sequence could be output in a burst. The input andcalculations can be continuous and require four clock cycles for everytwo bits of data in one non-limiting example. An input FIFO can bedependent on a depuncture input data rate.

It should also be understood that the present invention is not limitedto convolutional codes and similar FEC, but also turbo codes could beused as high-performance error correction codes or low-densityparity-check codes that approach the Shannon limit as the theoreticallimit of maximum information transfer rate over a noisy channel. Thus,some available bandwidth can be increased without increasing the powerof the transmission. Instead of producing binary digits from the signal,the front-end of the decoder could be designed to produce a likelihoodmeasure for each bit.

The system and extended preamble, in accordance with non-limitingexamples of the present invention, can be used in multiprocessorembedded systems and related methods and also used for any type of radiosoftware communications architecture as used on mainframe computers orsmall computers, including laptops with an added transceiver, such asused by military and civilian applications, or in a portable wirelesscommunications device. The portable wireless communications device caninclude a transceiver as an internal component and handheld housing withan antenna and control knobs. A Liquid Crystal Display (LCD) or similardisplay can be positioned on the housing in an appropriate location fordisplay. The various internal components, including dual processorsystems for red and black subsystems and software that is conformingwith SCA, can be operative with the radio. The architecture as describedcan be used with any processor system operative with the transceiver inaccordance with non-limiting examples of the present invention. Anexample of a communications device that could incorporate the system andmethod, in accordance with non-limiting examples of the presentinvention, is the Falcon® III manpack or tactical radio platformmanufactured by Harris Corporation of Melbourne, Fla.

This application is related to copending patent applications entitled,“COMMUNICATIONS DEVICE AND RELATED METHOD WITH REDUCED FALSE DETECTSDURING START OF MESSAGE BIT CORRELATION,” and “COMMUNICATIONS DEVICE ANDRELATED METHOD WITH IMPROVED ACQUISITION ESTIMATES OF FREQUENCY OFFSETAND PHASE ERROR,” which are filed on the same date and by the sameassignee and inventors, the disclosures which are hereby incorporated byreference.

Many modifications and other embodiments of the invention will come tothe mind of one skilled in the art having the benefit of the teachingspresented in the foregoing descriptions and the associated drawings.Therefore, it is understood that the invention is not to be limited tothe specific embodiments disclosed, and that modifications andembodiments are intended to be included within the scope of the appendedclaims.

1. A method of processing a communications signal, comprising: receivingwithin a modem a repeated preamble bit or symbol pattern for a digitallymodulated communications signal; generating an initial frequency offsetestimate and phase error estimate by detecting the repeated preamblepattern for a block of samples within the communications signal;correlating two halves of the block of samples with a plurality ofdifferent shifted sequences and determining the shifted sequence thatprovides the maximum correlation value; and establishing a symbol timingestimate based on the known timing alignment of the shifted sequencethat provides the maximum correlation value.
 2. The method according toclaim 1, which further comprises rotating the block of samples by thenegative of the initial frequency offset estimate, in order tocompensate for the received frequency offset.
 3. The method according toclaim 1, wherein the initial signal detection spans 256 samplescontaining 64 symbols sampled at 4 samples per symbol.
 4. The methodaccording to claim 1, which further comprises calculating themagnitude-squared of the sum of the two halves of the correlationoutputs for the plurality of shifted sequences to obtain a symbol timingalignment.
 5. The method according to claim 1, wherein the repeatedpreamble pattern comprises a training sequence.
 6. The method accordingto claim 1, and further comprising transmitting a start of messagesequence after the repeated preamble pattern.
 7. The method according toclaim 1, and further comprising transmitting a 011 pattern as therepeated preamble pattern.
 8. The method according to claim 1, whichfurther comprises correlating the two halves with 12 different shiftedsequences.
 9. A method of processing a communications signal,comprising: receiving within a modem a repeated preamble pattern as atraining sequence for a binary phase shift keyed (BPSK) communicationssignal; generating an initial frequency offset estimate and phase errorestimate by processing a Fast Fourier Transform (FFT) that spans a blockof samples for the BPSK communications signal and detects the repeatedpreamble bit pattern for the block of samples within the communicationssignal; rotating the block of samples by the negative of the initialfrequency offset estimate, in order to compensate for the receivedfrequency offset; correlating two halves of the block of samples with aplurality of different BPSK shifted sequences and determining theshifted sequence that provided the maximum correlation value; andestablishing a symbol timing estimate based on the known timingalignment of the shifted sequence that provided the maximum correlationvalue.
 10. The method according to claim 9, wherein the FFT spans 256samples containing 64 BPSK symbols sampled at 4 samples per symbol. 11.The method according to claim 9, which further comprises calculating themagnitude-squared of the sum of the two halves of the differentcorrelation outputs for the plurality of shifted sequences to obtain atiming alignment.
 12. The method according to claim 9, and furthercomprising transmitting a start of message sequence after the repeatedpreamble pattern.
 13. The method according to claim 9, and furthercomprising transmitting a 011 pattern as the repeated preamble pattern.14. The method according to claim 9, which further comprises updating afrequency offset estimate iteratively an N number of times using theshifted sequence that provided the maximum correlation value to refinean acquisition estimate of the frequency offset and phase error of thereceived communications signal.
 15. The method according to claim 9,which further comprises correlating the two halves with 12 differentBPSK shifted sequences.
 16. The method according to claim 9, and furthercomprising transmitting a start of message sequence after the repeatedpreamble pattern.
 17. The method according to claim 9, and furthercomprising transmitting a 011 pattern as the repeated preamble pattern.18. A communications device, comprising: a signal input for receiving abinary phase shift keyed (BPSK) communications signal having a repeatedpreamble pattern; a modem that processes the BPSK communications signaland further comprising a demodulator and processor that generates aninitial frequency offset estimate and phase error estimate by detectingthe repeated preamble bit pattern for a block of samples within thecommunications signal, correlates two halves of the block of sampleswith a plurality of different BPSK shifted sequences and determines theshifted sequence providing the maximum correlation value and uses theknown timing alignment of this sequence to establish a symbol timingestimate; and radio circuitry operative with the modem for processingcommunications data obtained from the communications signal.
 19. Thecommunications device according to claim 18, wherein said modem isoperative for rotating the block of samples by the negative of theinitial frequency offset estimate, in order to compensate for thereceived frequency offset.
 20. The communications device according toclaim 18, wherein said modem is operative for calculating themagnitude-squared of the sum of the two halves of the correlationoutputs for the plurality of shifted sequences to obtain a symbol timingestimate based on the shifted sequence providing the maximum correlationvalue.
 21. The communications device according to claim 18, wherein therepeated preamble pattern comprises a training sequence.
 22. Thecommunications device according to claim 18, wherein the initialdetection circuit spans 256 samples containing 64 BPSK symbols sampledat 4 samples per symbol.